Multiple-input-multiple-output transmission using non-binary LDPC coding

ABSTRACT

A wireless communication system constructed by a MIMO antenna system and transmitting information from a transmitter having Nt number of transmitting antennas to a receiver  3  having Nr number of receiving antennas. The receiver  3  of this wireless communication system  1  linking the inputs and outputs between a demodulating unit  32  demodulating the input signals from receiving antennas  31  and a decoding unit  35  receiving as input that demodulation output and decoding the non-binary LDPC code of the GF(q) one-to-one under predetermined conditions to enable the inputs and outputs of this demodulating unit  32  and decoding unit  35  to be directly connected and thereby improve the decoding characteristic. That predetermined condition is preferably q=2 Nt·m .

TECHNICAL FIELD

The present invention relates to a wireless communication system, inparticular a wireless communication system employing a multi-inputmulti-output (MIMO) antenna system, further relates to a method ofconstructing that wireless communication system and a receiver formingthe receiving side of that wireless communication system.

BACKGROUND ART

In recent years, the MIMO antenna system has come under attention as atransmission technology for wireless communications. This MIMO antennasystem provides a plurality of transmitting/receiving antennas at atransmitting side of a wireless communication system and provides aplurality of transmitting/receiving antennas at that receiving side andhas these multi-antennas transmit and receive information. In this MIMOantenna system, a large number of communication paths are formed betweenall antennas of the transmitting side and all antennas of the receivingside. As a result, the speed of data transmitted from the transmittingside to the receiving side is multiplied. For example, if using twotransmitting antennas, the speed of that transmitted data is doubled,while if using four, it is quadrupled. Therefore, the channel capacitybetween the transmitter and receiver can be greatly increased. However,in this case, it is assumed that the bit correlation between theplurality of transmitting antennas is sufficiently small and the bitcorrelation between the plurality of receiving antennas is sufficientlysmall.

On the other hand, further, in recent years, a LDPC (Low Density ParityCheck) code has begun to be used for encoding the transmitted data in awireless communication system etc. This LDPC code, like the conventionalturbo code, is an error correction code having superior characteristicsclose to the Shannon limit.

In particular, it is known that this LDPC code has a decodingcharacteristic equal to the turbo code or, when that code length islong, exhibits better characteristics than the turbo code. For example,when the code length is tens of thousands of bits or more, sometimes itexceeds the decoding characteristic of the turbo code currently employedin third generation mobile phone systems.

As the LDPC code, currently a binary type LDPC code and a non-binarytype LDPC code are known, but if the latter non-binary LDPC code isused, there is the defect that the amount of processing increasescompared with the former binary LDPC code. However, while there is sucha defect, if this non-binary LDPC code is used, even if the code lengthbecomes short, an improvement of the decoding characteristic can beexpected compared with use of the binary LDPC code.

Therefore, it can be easily understood that by using both the MIMOantenna system and non-binary LDPC coding system, a wirelesscommunication system can be realized which increases the channelcapacity between the transmitter and receiver and further simultaneouslyimproves the decoding characteristic.

When realizing a high efficiency wireless communication system usingthis MIMO antenna system and non-binary LDPC coding system, first, it isproposed, based on the general technique of constructing currentwireless communication systems using the general binary LDPC codingsystem for that MIMO antenna system, to construct a high efficiencywireless communication system using the intended non-binary LDPC codingsystem of the present invention together with that MIMO antenna system.

That is, a new technique utilizing the normal technique of using both aMIMO antenna system and general binary LDPC coding system realizes thehigh efficiency wireless communication system using both a MIMO antennasystem and non-binary LDPC coding system.

However, if trying to use the normal technique as it is to realize thehigh efficiency wireless communication system using both the non-binaryLDPC coding system and MIMO antenna system, due to the later explainedreasons, the problem arises of both the information regarding the bitcorrelation between the plurality of transmitting antennas in the MIMOantenna system and information regarding the inter-bit correlation inthe modulated symbols being lost (loss of correlation information) andof the decoding characteristic of the non-binary LDPC code at thereceiving side ending up deteriorating.

This being so, as already explained, the advantage of improvement of thedecoding characteristic expected when using a non-binary LDPC code endsup being cancelled out by the disadvantage of the deterioration of thedecoding characteristic due to the loss of correlation information. As aresult, the advantage of using the non-binary LDPC code is lost andtherefore realization of an effective high efficiency wirelesscommunication system becomes difficult.

Note that as known in prior art relating to the present invention, thereare the following [Non-Patent Document 1] and [Non-Patent Document 2].

-   [Non-Patent Document 1] M. C. Davey and D. MacKay, “Low-Density    Parity Check Codes over GF (q)”, IEEE Comm. Lett., Vol. 2, No. 6,    June 1998.-   [Non-Patent Document 2] F. Guo and L. Hanzo, “Low complexity    non-binary LDPC and modulation schemes over MIMO channels”,    Vehicular Technology Conference, 2004, VTC2004-Fall, vol. 2, pp.    1294-1298, September 2004.

DISCLOSURE OF THE INVENTION Problem to be Solved by the Invention

Therefore, the present invention, in view of the problems, has as itsobject to provide a dual “MIMO+non-binary LDPC” type wirelesscommunication system enabling transmission and reception of data withoutdeterioration of the decoding characteristic of the non-binary LDPC codeat the receiving side.

Further, it has as its object the provision of a method of constructingthat wireless communication system and a receiver forming the receivingside of a wireless communication system.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a view of an example of a transmitter forming a transmittingside of a wireless communication system according to the presentinvention,

FIG. 2 is a view of a first embodiment of a receiver forming a receivingside of a wireless communication system according to the presentinvention,

FIG. 3 is a view of the configuration of an SISO type general receiverin a wireless communication system using a non-binary decoding system,

FIG. 4 is a view of a known Tanner graph applied to the parity checkmatrix H of FIG. 5,

FIG. 5 is a view of a parity check matrix H as an example,

FIG. 6 is a graph of the results of comparison of decodingcharacteristics between the present invention and the prior art,

FIG. 7 is a view of a second embodiment of a receiver forming areceiving side of a wireless communication system according to thepresent invention,

FIG. 8 is a view of an example of a wireless communication systemprovided with an adaptive modulation function, and

FIG. 9 is a view of an example of a transmitter of a multi-beam MIMOantenna system.

MODE OF WORKING THE INVENTION

FIG. 1 is a view of an example of a transmitter forming a transmittingside of a wireless communication system according to the presentinvention. In the figure, reference number 1 indicates a wirelesscommunication system as a whole, while 2 indicates a transmitter formingthe transmitting side of the system 1. Note that a receiver 3 formingthe receiving side of the system 1 is shown in the later explained FIG.2, but the transmitter 2 also has the function of the receiver of FIG.2, and the receiver 3 also has the function of the transmitter ofFIG. 1. For example, one side of the system 1 forms a base station,while the other side forms a mobile station in this configuration.

In FIG. 1, the transmitter 2 has an LDPC encoder 22 receiving as inputinformation bits Ib to be encoded and encoding them by a non-binary LDPCcode and a modulation unit 25 modulating the encoded information bits Ipproduced at this LDPC encoder 22 and transmitting them to the receiver 2side through a wireless channel 27.

Here, that non-binary LDPC code will be explained.

The LDPC code having the coding rate R and the code length N is definedas a linear block code by a parity check matrix H of M rows and Ncolumns. This LDPC code is called a “non-binary LDPC code” when thatparity check matrix H is comprised of non-binary Galois Field elements.

Here, if expressing a Galois Field having q number of elements as GF(q),in the case of a non-binary LDPC code, q>2. In particular, a case whereq is a power of 2 is generally used. For differentiation, an LDPC codewhen q=2 is called a “binary LDPC code”. An LDPC code is defined as Hc=0using the parity check matrix H. The computation is performed on theGF(q). Here, the c is a code word vector of N rows and 1 column. Thatis, the vectors c satisfying the above equation Hc=0 for a certainparity check matrix H are defined as code words of the LDPC code. Theparity check matrix H of this LDPC code is characterized by a smalldensity of non-zero components.

On the other hand, an LDPC code is decoded on a Tanner graphcorresponding to the parity check matrix H (see the later explained FIG.4). This Tanner graph is comprised of N number of variable nodes and Mnumber of check nodes (see N and M of FIG. 5) and WcN=WrM number ofedges. These correspond to the columns and rows in the parity checkmatrix H and non-zero components. Here, We expresses column weighting(number of non-zero components present in column), while Wr expressesrow weighting (number of non-zero components present in row). Note thathere, for simplification, the case where the column weight and rowweight are constant (regular LDPC code) is considered.

For the decoding, a BP (Belief Propagation) algorithm on the Tannergraph is used. This is a type of repetitive decoding and switchesinformation (messages) between variable nodes and check nodes to makethe likelihoods of the bits converge to the sub-optimal value. Thereceiving characteristic of the LDPC code greatly depends on this Tannergraph, that is, the parity check matrix H.

Returning again to FIG. 1, the transmitter 2 is provided with an inputside converter 21 provided at the input side of the above-mentioned LDPCencoder 22 for converting the binary information bits Ib to thenon-binary multi-value symbols and an output side converter 23 providedat the output side of the LDPC encoder 22 for converting the encodedsymbols of the non-binary multi-value symbols encoded by this LDPCencoder 22 to binary bits.

The input side converter 21 performs the multi-value conversion of,taking the example of the case of GF(4),

00→0

01→1

10→2

11→3

(GF(2)→GF(4)) for matching the binary information bits Ib to themulti-value being operated by non-binary LDPC encoder 22, while

the output side converter 23 similarly performs the binary conversionof, taking as an example the case of GF(4),

0→00

1→01

2→10

3→11

(GF(4)→GF(2)), for binary operation of the multi-value output from theLDPC encoder 22.

The information bits from the non-binary LDPC encoder 22 accompanyingthe operations of the input side converter (GF(2)→GF(q)) 21 and outputside converter (GF(q)→GF(2)) 23, as already explained, are modulated bythe modulation unit 25, then transmitted over the wireless channel 27.In this case, the wireless communication system 1 according to thepresent invention applies the MIMO antenna system to the above-mentionednon-binary LDPC coding system and further has a MIMO configurationcompatible with this MIMO antenna system.

That is, for this MIMO configuration, first the information bits fromthe non-binary encoders (21,22,23) are converted from a serial toparallel (S/P) format. This is performed by the serial/parallel (S/P)converter 24. The plurality (Nt) of parallel modulated symbols outputfrom this are transmitted over the wireless channel 27 from thecorresponding antennas of the Nt number of individual transmittingantennas 26-0 to 26-(Nt−1) at the transmitting multi-antenna part 26.

Therefore, the above-mentioned modulation unit 25 is further comprisedof a plurality (Nt) of individual modulation units 25-0 to 25-(Nt−1)corresponding to the individual transmitting antennas 26-0 to 26-(Nt−1).Here, the individual modulation units (25-0 to 25-(Nt−1)) receive asinput and modulate the parallel bits comprised of groups of m bits.Here, m is the number of bits per modulated symbol, that is, the orderof modulation. For example, if the transmitter 2 employs 16 QAM(Quadrature Amplitude Modulation) as the modulation scheme, m=4, whileif 64 QAM, m=6. Further, if employing QPSK (Quadrature Phase ShiftKeying), m=2. This order of modulation m, as explained later, becomes animportant parameter together with the number Nt of the individualtransmitting antennas (26-0 to 26-(Nt−1)) in constructing the wirelesscommunication system 1 according to the present invention.

In the above, the transmitter 2 was explained. Next, the receiver 3 ofFIG. 2 according to the present invention will be explained. Beforethat, first, referring to FIG. 3, in general, that is, not a MIMO type,but an SISO (Single Input Single Output) type receiver, employing ageneral non-binary reception processor will be explained as the receiver4.

In this FIG. 3, the receiver 4 is a receiver including a demodulatingunit 42 receiving a transmission signal encoded by a non-binary LDPCcode (22), modulated (25), and transmitted by the transmitting side (2)and demodulating it and an LDPC decoder 44 decoding the demodulated bitsexpressed by the bit likelihood from this demodulating unit 42 by anon-binary LDPC code.

This receiver 4 further is provided with an input side converter 43provided at the input side of the non-binary LDPC decoder 44 forconverting the binary demodulated bits expressed by the bit likelihoodfrom the demodulating unit 42 to non-binary multi-value symbols and anoutput side converter 45 provided at the output side of the LDPC decoder44 for converting the information bits comprised of the non-binarymulti-value symbols decoded by that LDPC decoder 44 to binaryinformation bits Ib.

That is, this output side converter 45 performs the binary conversion of

0→00

1→01

2→10

3→11

(GF(4)→GF(2)).

On the other hand, the input side converter 43 is suitably called a “bitlikelihood converter” judging from that function. This converter groupstogether the binary (bit) likelihoods obtained by the demodulating unit42 for each p (=log₂(q)) bits and converts them to one symbol ofnon-binary likelihood. As an example, the GF(2)→GF(4) conversionbecomes:

Q₀=P₁₀P₂₀

Q₁=P₁₀P₂₁

Q₂=P₁₁P₂₀

Q₃=P₁₁P₂₁

Here, P₁₀ is the probability of the 2nd bit in the two bits to beconverted to a GF(4) symbol being 0, while Q₀ is the probability of theGF(4) symbol being 0.

The thus obtained symbol likelihood is input to the LDPC decoder 44.This decoder 44 decodes the LDPC code by the already explained Tannergraph (FIG. 4).

FIG. 4 is a view showing a Tanner graph corresponding to the paritycheck matrix H of FIG. 5, FIG. 5 is a view of a parity check matrix Hgiven as an example.

In FIG. 4, V₀, V₁, . . . V₅ are so-called variable nodes, while C₀, C₁ .. . C₃ are so-called check nodes. The F₀, F₁ . . . F₅ input to thevariable nodes (V) are the results of demodulation from the demodulatingunit 42 of FIG. 3 through the above-mentioned bit likelihood converter(43). Note that the bold line CL6 in FIG. 4 corresponds to the cycle CLshown in FIG. 5. This length is, in the illustrated example, 6.

Here, returning to FIG. 2, this FIG. 2 is a view of a first embodimentof the receiver 3 forming the receiving side of the wirelesscommunication system 1 according to the present invention. Note that inthis wireless communication system 1, the plurality (Nt) of individualtransmitting antennas 26-0 to 26-(Nt−1) shown in FIG. 1 simultaneouslytransmit different signals in the same frequency band.

On the other hand, the receiver 3 of FIG. 2 employs a MIMO antennasystem receiving the composite transmission signal transmitted in thesame band in the same time through the wireless channel 27 by thereceiving multi-antenna part 31 comprised of the plurality (Nr) ofseparate receiving antennas 31-0 to 31-(Nr−1) and performingpredetermined signal reception processing to separate the combinedtransmission signal. Note that the number of transmitting antennas (26)and the number of receiving antennas (31) do not necessarily have to bethe same, so these are expressed by the above Nt and Nr.

As shown in FIG. 2, the receiver 3 according to the present inventionreceives the combined transmission signal as an input signal at thereceiving multi-antenna part 31, the demodulating unit 32 comprised ofthe channel coefficient estimating unit 33 and demodulator 34demodulates this, and the non-binary LDPC decoding unit 35 decodes thedemodulated bits. The decoded symbols of dimension (q) from thisdecoding unit (GF(q)) 35 are passed through the output side converter 36(same as 45 of FIG. 3) converting it to binary bits of dimension (2) tobe reproduced as the original information bits Ib.

Here, in FIG. 2, taking note of the demodulating unit 32 and itsperiphery characterizing the present invention, first, in theabove-mentioned FIG. 1, the binary transmission bit series of length pK(p=log₂q) of dimension (2) is converted to a series of symbol of lengthK comprised of dimension (q) expressed by the number of elements q.Further, as already explained, that series of symbol is encoded by thenon-binary LDPC encoder 22 of dimension (q), then a series of encodedsymbols of length N is produced, then is converted by the converter 23again to the dimension (2), then is modulated by the modulation unit(25-0 to 25-(Nt−1)) for each of the individual transmitting antennas(26-0 to 26-(Nt−1)) and transmitted to the receiver 3 side of FIG. 2.

The receiver 3 of FIG. 2 estimates the complex wireless channelcoefficients between the individual antennas 31-0 to 31-(Nr−1) and theindividual transmitting antennas 26-0 to 26-(Nt−1). This is performed bythe channel coefficient estimating unit 33. Note that this channelcoefficient estimation can be performed by using for example orthogonalknown pilot signals between individual transmitting antennas.

Using this estimated channel coefficient matrix as Ω, the demodulator 34uses the later explained predetermined relation (1) to calculate theinput signal (demodulated bits) to the non-binary LDPC decoding unit 35.Based on this input signal, the decoding unit 35 performs non-binarydecoding and further the dimension converting unit 36 converts theresult to a binary system to finally obtain a binary received bit series(information bits Ib).

Here, the points of the receiver 3 shown in FIG. 2 will be summarized.First, the receiver 3 predicated on is a receiver provided with aplurality of receiving antennas 31 for transmitting and receivinginformation with a transmitter 2 provided with a plurality oftransmitting antennas 26 under a multi-input multi-output (MIMO) antennasystem and uses a non-binary LDPC code as the channel coding scheme fortransmission/reception with that transmitter 2.

Here, this receiver 3 is a receiver at a receiving part including ademodulating unit 32 demodulating input signal information from areceiving antennas 31 and a decoding unit 35 decoding that input signalin accordance with the non-binary LDPC code at the output side of thisdemodulating unit 32, which receiver can make the output group ofmulti-value symbols expressed by the symbol likelihood from thatdemodulating unit 32 and the input group of the decoding unit 35comprised of the non-binary LDPC code of the predetermined GF(q) matchone-to-one to directly connect these demodulating unit 32 and decodingunit 35.

That is, if assuming that for example 256 types of symbol likelihood areoutput from the demodulator 34 of the demodulating unit 32 (at j of FIG.2, the group of outputs is “0, 1 . . . 255”), the non-binary LDPCdecoding unit 35 is made of a GF (256) LDPC decoding unit and that groupof 256 types of input are linked completely one-to-one with the group of256 types of outputs “0, 1 . . . 255”.

Conversely, if the non-binary LDPC decoding unit 35 is comprised of GF(256), a wireless communication system 1 is configured where thedemodulator 34 gives 256 types of output (symbol likelihoods).

In this way, the outputs of the demodulator 34 at the receiver 3 and theinputs of the decoding unit 35 are made to completely match one-to-one,so the later explained conventional marginalization can be eliminated.That is, in a wireless communication system 1 including this receiver 3,there is no longer the past problem of (i) the information on the bitcorrelation between the plurality of transmitting antennas and (ii) theinformation on the inter-bit correlation in the modulated symbols beinglost before the decoding. Therefore, there is no longer the problem ofthe decoding characteristic of the non-binary LDPC code at the receiver3 deteriorating.

Summarizing the points of the wireless communication system 1 accordingto the present invention in the same way as the above points of thereceiver 3, first, the predicated wireless communication system is awireless communication system comprised of a transmitter 2 provided witha plurality of transmitting antennas 26 and a receiver 3 provided with aplurality of receiving antennas 31 transmitting and receivinginformation under a multi-input multi-output (MIMO) antenna system andusing a non-binary LDPC code as the channel coding scheme between thesetransmitter 2 and receiver 3. Here, the above receiver 3, which includesa demodulating unit 32 demodulating an input signal from a receivingantenna 26 and a decoding unit 35 at the output side of thisdemodulating unit 32 for decoding the input signal in accordance withthe non-binary LDPC code is configured so as to make the group ofoutputs of multi-value symbols expressed by symbol likelihood from thedemodulating unit 32 and the group of inputs of the decoding unit 35comprised of the GF(q) non-binary LDPC code match one-to-one to enablethe demodulating unit and decoding unit to be directly connected.

As explained above, in the receiver 3 in the wireless communicationsystem 1 according to the present invention, the point is to make theoutput of the demodulator 34 and the input of the decoding unit 35completely match one-to-one. To realize such a one-to-one completematch, a predetermined condition must be satisfied.

For this predetermined condition, the number of transmitting antennas,order of modulation, and dimensions q of GF(q) (number of elements) areimportant parameters.

More specifically, the number (Nt) of the plurality of transmittingantennas 26-0 to 26-(Nt−1) and the order of modulation (m) of the numberof bits per modulated symbol forming the signal modulated by apredetermined modulation scheme and transmitted from the transmitter 2are set linked with the dimension q of GF(q).

More specifically, the above-mentioned number (Nt) of the plurality oftransmitting antennas and the order of modulation (m) may be set bylinking them with the dimension q of GF(q) based on the followingcondition equation (1):q=2^(Nt·m)  (1)

As a general example, assume that (i) the transmitting multi-antennapart 26 is comprised of two individual transmitting antennas 26-0 and26-1 and (ii) the transmission signal modulated by 16 QAM from thetransmitter 2 is transmitted from that antenna part 26 toward thereceiver 3. That is, for the (i), the number Nt of antennas becomes 2,while for the (ii), the order of modulation m becomes 4.

This being so, the dimension q of a GF(q) is q=2^(2·4)(=2⁸), that is,256. That is, in a wireless communication system 1 where the values Nt=2and m=4, by making the dimension of the non-binary LDPC decoding unit 35“256”, as already explained, the decoding unit 35 performs decodingwithout loss of (i) the information regarding the bit correlation amongthe plurality of transmitting antennas and (ii) the informationregarding the bit correlation in the modulated symbols before thedecoding. Therefore, compared with a conventional receiver 4 accompaniedwith loss of these information, the deterioration of the decodingcharacteristic is greatly reduced.

As explained above, under the condition of q=2^(Nt·m), when the outputof the demodulating unit 32 and the input of the decoding unit 35 aremade to completely match one-to-one, the following condition equation(2) of F_(k) ^(j) stands. Conversely speaking, if designing thedemodulator 34 of the demodulating unit 32 so as to determine the symbollikelihood (F) satisfying the condition equation (2) of F_(k) ^(j),one-to-one linkage with the decoding unit 35 becomes possible. Thatcondition equation (2) is the following

$\begin{matrix}{F_{k}^{j} = {\alpha\;{\exp\left\lbrack {- \frac{{{y_{k} - {\Omega\; x_{j}}}}^{2}}{\sigma^{2}}} \right\rbrack}}} & (2)\end{matrix}$Here, the index (see “j” between the demodulator 34 and decoding unit 35of FIG. 2) expresses the order (0, 1 . . . , q−1) assigned to the groupof q number of inputs (symbol likelihoods), while index “k” expressesthe time index of the input of the symbol likelihood. Further, “y_(k)”expresses a k-th received signal vector, “x_(j)” expresses a replica forthe j-th transmission signal vector (explained later), and “Ω”multiplied with that x_(j) expresses a channel coefficient matrixestimated by the channel coefficient estimating unit 33. On the otherhand, the denominator side “σ²” expresses an average power of noise,further the exp coefficient “α” expresses a normalization constant fornormalization so that a sum of F_(k) ^(j) relating to all j becomes 1.Further, “∥ ∥²” (norm) at the numerator expresses a sum of the square ofthe absolute value of each element in the vector.

Here it should be noted that the equation of the F_(k) ³ does notinclude the symbol Σ (symbol of addition). This addition operation isessential in the above-mentioned “marginalization” of the prior art, butaccording to the present invention, this “marginalization” iseliminated. Therefore, the problem of the above-mentioned “loss ofcorrelation information” due to this marginalization disappears. Thatis, the deterioration of the decoding characteristic at the receiverside is greatly reduced. That effect is shown by a graph.

FIG. 6 is a graph showing the results of comparison of the decodingcharacteristics between the present invention and the prior art. Thatis, it is a graph showing an example of the results of simulation forclarifying the excellent effects according to the present invention.

In the graph of FIG. 6, the ordinate shows the “block error rate”(BLER), while the abscissa shows the “ratio of the signal power per bitto the density of the noise power (Eb/No) [dB]”. Further, as shown atthe top right of the graph, it is an example of the case of using noisecomprised of Additive White Gaussian Noise (AWGN), employing a twotransmitting antenna/two receiving antenna MIMO antenna system, andusing a 16 QAM modulation scheme.

In the graph, the curve “a” shows the characteristic of the case of theconventional general binary LDPC coding scheme, while the curve “b”shows the characteristic of the case of the normal non-binary LDPCcoding scheme accompanied with the above-mentioned“marginalization”—both conventional types.

As opposed to this, the curve “c” shows the characteristic in the caseof demodulation based on the non-binary LDPC coding/decoding schemeaccording to the present invention, that is, the equation (2).

Comparing these curves “a”, “b”, and “c”, it is learned that the presentinvention (“c”) greatly improves the BLER compared with the conventionaltypes (“a”, “b”) and has almost none of the already explained“deterioration of the decoding characteristic”.

Here, the “marginalization” causing the already explained “deteriorationof the decoding characteristic” will be explained. In general, as thesignal processing at the receiving side of a wireless communicationsystem, MMSE (Minimum Mean Square Error) demodulation (equalization) orMLD (Maximum Probability Detection) demodulation (equalization) is used.

This MLD demodulation is a scheme using “replicas” of all possibletransmitted bit patterns to find the post-establishment of the bits.“Replica” means the data D′ which should be received at the receivingside when the transmitting side transmits the data D. For example, inthe case of the 4-bit data D, 16 replicas (D′) are formed.

The MLD demodulation naturally involves a larger amount of processingcompared with the MMSE demodulation of just linear processing, butconversely use of this greatly improves the decoding characteristic.

Here, if the channel coefficient matrix, as explained regarding thechannel coefficient estimating unit 33 of FIG. 2, is Ω, the receivedsignal vector from the receiving antenna 31 is y, and the noise vectoris n, the function among these is expressed by the following equation:y=Ωx+n  (4)At this time, the MMSE weight is given byW=Ω ^(H)(ΩΩ^(H)+σ² I)⁻¹  (5)Here, σ² is the average noise power shown by the above equation (2),while I is the unit matrix. This MMSE demodulation (equalization) isobtained by multiplying the received signal vector y with the MMSEweight W.

The MLD demodulation giving a better decoding characteristic than thisMMSE demodulation finds the already explained bit likelihood (F) inaccordance with the following equation (6).

$\begin{matrix}{F_{j} = \frac{\sum\limits_{{x:{bj}} = 0}{\exp\left\lbrack {- \frac{{{y - {\Omega\; x}}}^{2}}{\sigma^{2}}} \right\rbrack}}{\sum\limits_{{x:{bj}} = 1}{\exp\left\lbrack {- \frac{{{y - {\Omega\; x}}}^{2}}{\sigma^{2}}} \right\rbrack}}} & (6)\end{matrix}$

At the numerator of the equation (6),

$\sum\limits_{{x:{bj}} = 0}$means to add all of the replicas where the j-th (for example, when x is4 bits, j=1, 2, 3 . . . 16) bit in the binary pattern of “1” and “0” ofall of the above-mentioned transmission bits x is “0” to find the2^(Nt·m−1) number of sums,

while at the denominator of the equation (6), means to add all of thereplicas where the j-th bit in the binary pattern of “1” and “0” of allof the above-mentioned transmission bits x is “1” to find the2^(Nt·m−1)number of sums

$\sum\limits_{{x:{bj}} = 1}.$

The bit likelihood (F) is based on the known equationF=Pr(b=0|r)/Pr(b=1|r)whereby the equation (6) is obtained. Here, Pr(b=0|r) means theprobability that a modulated and then transmitted bit b is “0” under thecondition that the received signal is r,

while Pr(b=1|r) means the probability that a modulated and thentransmitted bit b is “1” under the condition that the received signal isr.

The MLD demodulation was explained predicated on use of a binary LDPCcode. When using a non-binary code upon which the present invention ispredicated, the bit probabilities are combined, the symbol likelihood isrecalculated, then this is used as the input of the decoding unit 35.

In the end, if adopting the conventional method of using the MLDdemodulation for calculation of the symbol likelihood, the routinestarts by finding first the bit likelihood. This being so, forcalculation of this bit likelihood, an operation for addition by E ofthe equation (6), that is, “marginalization”, becomes essential and theabove-mentioned “loss of correlation information” is caused. Due tothis, deterioration of the already explained decoding characteristic isinvited.

On the other hand, according to the equation (2) forming the basis ofthe routine for calculation of the symbol likelihood based on thepresent invention, there is no operation of addition by theabove-mentioned E. The q number of symbol likelihoods obtained by thisequation (2) can be directly connected as is to the q number of inputsof the decoding unit 35 of the GF(q).

Next, a second embodiment according to the present invention and amethod of configuration of a wireless communication system will beexplained.

FIG. 7 is a view showing a second embodiment of a receiver 3 forming thereceiving side of the wireless communication system 1 according to thepresent invention. This second embodiment is characterized by linkingthe number (Nt) of the plurality of transmitting antennas and the orderof modulation (m) with the dimension q of the GF(q) based on thefollowing condition equation (3):q=2^(Nt·m·m)

(where, n is any natural number)

Specifically, a dimension converting unit 37 receiving the q′(=2^(Nt·m)) number of outputs from the demodulating unit 32 as input andinputting this as q (=2^(Nt·m·n)) number of outputs to the decoding unit35 is interposed between the demodulating unit 32 and decoding unit 35.Note that this second embodiment is exactly the same as theabove-mentioned first embodiment (FIG. 2) except for the introduction ofthat dimension converting unit 37.

In the case of this first embodiment, the demodulating unit 32 anddecoding unit 35 were directly connected completely one-to-one, butdepending on the system design of the wireless system 1, sometimes thedimension q′ of the output from the demodulating unit 32 does not matchthe dimension q of the GF(q) of the input at the decoding unit 35 andq>q′.

The second embodiment, in this case, can convert the q′ to q tosubstantively make the demodulating unit 32 and decoding unit 35correspond one-to-one to enable them to be directly connected. This isperformed by the dimension converting unit 37. As illustrated, itconverts the dimensions from GF(q′) to GF(q).

In this way, according to the second embodiment, there is the advantagethat the present invention can be applied even if the demodulating unit32 and decoding unit 35 do not match in dimension by just inserting thedimension converting unit 37, but there is also another advantage.

In general, a non-binary LDPC code increases the Galois Field dimensions(q). Along with this, the amount of processing increases, but thedecoding characteristic tends to become better. The second embodimentincreases the dimensions of the demodulating unit 32 by 2^(n) fold, sogives rise to the advantage of a much better decoding characteristic.

Next, a method of configuration of a wireless communication systemaccording to the present invention will be explained. This method ofconfiguration is a method of configuration of a wireless communicationsystem 1, as explained above, comprised of a transmitter 2 provided witha plurality of transmitting antennas 26 and a receiver 3 provided with aplurality of receiving antennas 31 transmitting and receivinginformation under a multi-input multi-output (MIMO) antenna system andusing a non-binary LDPC code as the channel coding scheme between thesetransmitter 2 and receiver 3, wherein this receiver 3 includes ademodulating unit 32 for demodulating the input signal from thereceiving antennas 31 and a decoding unit 35 at the output side of thisdemodulating unit 32 for decoding that input signal according to thenon-binary LDPC code. Here, the method of configuration has at least afirst step and second step.

At the first step, the dimension q of the non-binary LDPC code, linkedwith the number (Nt) of the plurality of transmitting antennas 26 andthe number of bits per modulated symbol forming the signal of theinformation demodulated by a predetermined modulation scheme andtransmitted from the transmitter 2, constituting the order of modulation(m), is set. Further,

at the second step, the group of outputs of the multi-value symbolsexpressed by the symbol likelihoods from the demodulating unit 32 andthe group of inputs of the decoding unit 35 comprised of the non-binaryLDPC code of dimensions (q) are made to match one-to-one and thesedemodulating unit 32 and decoding unit 35 are directly connected.

When the wireless communication system 1 is a wireless communicationsystem provided with an adaptive modulation function, the number oftransmitting antennas and modulation scheme frequently change accordingto the state of the channel coefficient (wireless channel 27). In thecase of this wireless communication system, the dimension (q) set at thefirst step can be suitably determined in accordance with the values ofthe number (Nt) of transmitting antennas and order of modulation (m)determined in accordance with the configuration of that wireless system.

FIG. 8 is a view of an example of a wireless communication systemprovided with an adaptive modulation function. This wirelesscommunication system 1 is comprised of for example a base station 51 anda mobile station 52 facing each other across a wireless channel 27.These stations 51 and 52 are provided with pluralities oftransmitting/receiving antennas 26/31.

In this wireless communication system of FIG. 8, before the first step,a notification step is provided. This notification step notifies thereceiver (mobile station 52) of transmission specification information(see FIG. 7) including at least the number (Nt) of transmitting antennasand order of modulation (m) determined in accordance with theconfiguration of the wireless system 1.

Note that the second step may include a step of reconstructing thedemodulating unit 32 so as to make the above-mentioned group of outputsand group of inputs match one-to-one or

this second step may include a step converting the dimensions betweenthe demodulating unit 32 and decoding unit 35 so as to make theabove-mentioned group of outputs and group of inputs match one-to-one(see dimension converter 37 of FIG. 7).

Referring to FIG. 8 again, when considering downlink adaptivemodulation, the mobile station 52 measures the state of the downlinkwireless channel and notifies it to the base station 51 through anuplink. The base station 51 determines the number (Nt) of transmittingantennas, the modulation scheme (m), the coding rate (R), etc. based onthe received downlink channel coefficient information T. Simultaneously,it determines the dimension (q) of the non-binary LDPC code used. In theabove preferred example, q=2^(Nt·m).

The base station 51 performs coding and modulation in accordance withthe determined parameters and transmits the transmission signal.Simultaneously, it notifies the mobile station 52 of these parameters(the transmission specification information S). The mobile station 52receiving the parameters (S) performs demodulation (32) and decoding(35) based on the parameters. In this case, as already explained, thedimension (q) of the non-binary LDPC code is uniquely determined fromthe information of the parameters (S) received.

The wireless communication system 1 of the present invention explainedabove may not only be configured by a conventional MIMO antenna system,but may also be configured by a “multi-beam MIMO” antenna systemproposed in 3GPP in recent years. FIG. 9 is a view of an example of atransmitter according to a multi-beam MIMO antenna system.

The component to note in FIG. 9 is a multi-beam generating unit 61. Thisforms a multi-beam MIMO. Therefore, the MIMO antenna system explainedwith reference to FIG. 1, FIG. 2, etc. may be made a multi-beam MIMOantenna system where the transmitter 2 as shown in FIG. 9 transmits thetransmission signals from the transmitting antennas multiplied with afixed weight.

As explained above, according to the present invention, as shown in FIG.6, a wireless communication system is realized which can greatlysuppress the “deterioration of the decoding characteristic” at thereceiver side.

1. A wireless communication system comprising a transmitter providedwith a plurality of transmitting antennas and a receiver provided with aplurality of receiving antennas transmitting and receiving informationunder a multi-input multi-output (MIMO) antenna system and using anon-binary LDPC code as a channel coding scheme between said transmitterand receiver, said receiver comprising: a demodulating unit fordemodulating an input signal from said receiving antennas and a decodingunit at the output side of said demodulating unit for decoding saidinput signal according with said non-binary LDPC code, wherein thereceiver makes a group of outputs of multi-value symbols expressed bysymbol likelihood, output from said demodulating unit and the group ofinputs of said decoding unit comprised of a non-binary LDPC code ofpredetermined dimension (q) match one-to-one to enable said demodulatingunit and said decoding unit to be directly connected, wherein a number(Nt) of said plurality of transmitting antennas and a number of bits permodulation signal forming a signal of said information modulated by apredetermined modulation scheme and transmitted from said transmitter,constituting an order of modulation (m), are set with said dimension (q)and said demodulating unit determining said symbol likelihood (F) inaccordance with the following condition equation:$F_{k}^{j} = {\alpha\;{\exp\left\lbrack {- \frac{{{y_{k} - {\Omega\; x_{j}}}}^{2}}{\sigma^{2}}} \right\rbrack}}$(where, an index “j” expresses an order (0, 1 . . . q−1) assigned to agroup of q number of inputs, an index “k” expresses the order of inputof symbol likelihood in a time direction, “y_(k)” expresses a k-threceived signal vector, “x_(j)” expresses a replica for a j-thtransmission signal vector, “Ω” expresses a channel coefficient matrixestimated by channel coefficient estimation, “σ²” expresses an averagepower of noise, “α” expresses a normalization constant for normalizationso that a sum of F_(k) ^(j) relating to all j becomes 1, “∥ ∥²” (norm)expresses a sum of the square of the absolute value of each element inthe vector), and said number (Nt) of the plurality of transmittingantennas and said order of modulation (m) are linked with the dimension(q) based on the following condition equation:q=2^(Nt·m·n)(where, n is any positive integer).
 2. A wirelesscommunication system as set forth in claim 1, characterized in that adimension converting unit receiving as input q′ (=2^(Nt·m)) number ofoutputs from said demodulating unit and inputting q (=2^(Nt·m·n)) numberof outputs to said decoding unit is interposed between said demodulatingunit and decoding unit.
 3. A receiver provided with a plurality ofreceiving antennas for transmitting and receiving information with atransmitter provided with a plurality of transmitting antennas under amulti-input multi-output (MIMO) antenna system and using a non-binaryLDPC code as a channel coding scheme for transmission and reception withsaid transmitter, said receiver making a group of outputs of multi-valuesymbols expressed by symbol likelihood from a demodulating unit and agroup of inputs of a decoding unit comprised of a non-binary LDPC codeof a predetermined dimension (q) match one-to-one to enable saiddemodulating unit and said decoding unit to be directly connected in areceiving part including a demodulating unit demodulating informationinput from said receiving antennas and a decoding unit at an output sideof said demodulating unit and decoding said input signal in accordancewith said non-binary LDPC code, wherein a number (Nt) of said pluralityof transmitting antennas and a number of bits per modulation signalforming a signal of said information modulated by a predeterminedmodulation scheme and transmitted from said transmitter, constituting anorder of modulation (m), are set with said dimension (q) and saiddemodulating unit determining said symbol likelihood (F) in accordancewith the following condition equation:$F_{k}^{j} = {{\alpha exp}\left\lbrack \frac{{{y_{k} - {\Omega\; x_{j}}}}^{2}}{\sigma^{2}} \right\rbrack}$(where, an index “j” expresses an order (0, 1 . . . q−1) assigned to agroup of q number of inputs, an index “k” expresses an order of input ofsymbol likelihood in a time direction, “y_(k)” expresses a k-th receivedsignal vector, “x_(j)” expresses a replica for a j-th transmissionsignal vector, “Ω” expresses a channel coefficient matrix estimated bychannel coefficient estimation, “σ⁻²” expresses an average power ofnoise, “α” expresses a normalization constant for normalization so thata sum of F_(k) ^(j) relating to all j becomes 1, “∥ ∥²” (norm) expressesa sum of the square of the absolute value of each element in thevector), and said number (Nt) of the plurality of receiving antennas andsaid order of modulation (m) are linked with said (q) based on thefollowing condition equation:q=2^(Nt·m·n)(where, n is any natural number).
 4. A receiver as set forthin claim 3, characterized in that a dimension converting unit receivingas input q′ (=2^(Nt·m)) number of outputs from said demodulating unitand inputting q (=2^(Nt·m·n)) number of outputs to said decoding unit isinterposed between said demodulating unit and decoding unit.
 5. A methodof configuration of a wireless communication system comprised of atransmitter provided with a plurality of transmitting antennas and areceiver provided with a plurality of receiving antennas transmittingand receiving information under a multi-input multi-output (MIMO)antenna system and using a non-binary LDPC code as a channel codingscheme between said transmitter and receiver, said receiver, comprisinga demodulating unit demodulating information input from said receivingantennas and a decoding unit at an output side of said demodulating unitand decoding said input information by using said non-binary LDPC code,performs a first step, at the demodulating unit, of setting thedimension q of said non-binary LDPC code, linked with the number (Nt) ofsaid plurality of transmitting antennas and the number of bits permodulated symbol forming the signal of said information demodulated by apredetermined modulation scheme and transmitted from said transmitter,constituting the order of modulation (m), where the dimension q isadaptively determined in accordance with values of the number (Nt) ofsaid transmitting antennas and said order of modulation (m) determinedin accordance with the configuration of said wireless s stem and saidnumber Nt of the plurality of receiving antennas and said order ofmodulation (m) are linked with said (q) based on the following conditionequation:q=2^(Nt·m·n)(where, n is any natural number), and a second step ofmaking a group of outputs of multi-value symbols expressed by the symbollikelihoods from said demodulating unit and a group of inputs of thedecoding unit comprised of the non-binary LDPC code of dimension q matchone-to-one to directly connect these demodulating unit and decodingunit, wherein said demodulating unit determines said symbol likelihood(F) in accordance with the following condition equation:$F_{k}^{j} = {\alpha\;{\exp\left\lbrack {- \frac{{{y_{k} - {\Omega\; x_{j}}}}^{2}}{\sigma^{2}}} \right\rbrack}}$(where, an index “j” expresses an order (0,1 . . . q−1) assigned to agroup of q number of inputs, an index “k” expresses an order of input ofsymbol likelihood in a time direction, “y_(k)” expresses a k-th receivedsignal vector, “x_(j)” expresses a replica for a j-th transmissionsignal vector, “Ω” expresses a channel coefficient matrix estimated bychannel coefficient estimation, “δ²” expresses an average power ofnoise, “α” expresses a normalization constant for normalization so thata sum of F_(k) ^(j) relating to all j becomes 1, “∥ ∥²” (norm) expressesa sum of the square of the absolute value of each element in thevector).
 6. A method of configuration of wireless communication systemas set forth in claim 5, characterized by providing, before said firststep, a notification step of notifying said receiver of transmissionspecification information including at least the values of the number(Nt) of said transmitting antennas and said order of modulation (m)determined in accordance with the configuration of said wireless system.7. A method of configuration of wireless communication system as setforth in claim 5, characterized in that said second step includes a stepof reconstructing said demodulating unit to make said group of outputsand said group of inputs match one-to-one.
 8. A method of configurationof a wireless communication system as set forth in claim 5,characterized in that said second step includes a step of convertingdimensions between said demodulating unit and said decoding unit to makesaid group of outputs and said group of inputs match one-to-one.